In bioimpedance spectroscopy (BIS), the electric conductivity of a tissue is measured via electrodes, whose impedance depends on their geometry and composition . A known alternating current from a signal source is injected into the tissue under test and the resulting voltage is measured. To minimize the impact of the skin impedance, electrode impedance and the measurement system on the acquired data, often a tetra-polar measuring method is used, which requires a second set of electrodes to measure the voltage voltage drop on the bioimpedance [2, 3].
Due to its simplicity, BIS lays the foundation for more complex and medically relevant applications like electrical impedance tomography (EIT), as well as impedance pneumography and impedance cardiography. The former are used for harmless, real-time bedside pulmonary monitoring and are still subject to research .
The impedance of human tissue itself is frequency-dependent and also tissue-specific. Its characteristics in the frequency domain can be divided into three dispersions (
For a successful measurement, the signal of the source should not change the electrical characteristics of the probed tissue. Further, the injected signal has to be of low magnitude to prevent harm to the patient. Due to the need to inject low currents at low frequencies, galvanic coupled electrodes are used. Common mode signals easily evoke polarisation and therefore corrosion at the electrode skin interface [8, 9]. To prevent skin lesions, measurement artefacts and distortions, the common mode signal should be as low as possible .
In BIS, the source for the alternating excitation signal traditionally is a bidirectional “voltage-controlled current source” (VCCS). By design, the main advantage is the limited and adjustable current, with the result that maximum allowable current limits can inherently be achieved. Different current source topologies for floating and grounded loads exist like operational transconductance amplifier, mirrored current-conveyor [10, 11], load-in-the-loop, Tietze and Howland topologies .
Often, BIS applications use an Enhanced Howland Current Source (EHCS) with grounded load. Its simple layout with one op-amp and five matched resistors provide a constant current with a high output impedance independent of the load. Most simulated current sources achieve a high output impedance . Above 100 kHz, however, real implementations of the EHCS have stability problems caused by mismatching and tolerances of the resistors, stray-capacitance and frequency-dependent characteristics of the active elements. As a consequence, the output impedance drops at higher frequencies [3, 12] and also the common mode signal increases . In in-vivo measurements, the stability and performance of the EHCS continues to degrade further due to the capacitive configuration of the load caused by the electrode contact impedance, cable capacitance and the capacitance to ground .
A traditional EHCS for portable BIS applications by Xu et al. achieved an output impedance of 100 kΩup to 100 kHz . The proposed design became unstable for higher frequencies. The stability of EHCSs can be increased by compensation capacitors. For example, Nouri et al. achieved up to 2.8MΩat 1 MHz . Recently, Saulnier et al. presented an FPGA-based adaptive algorithm to adjust the EHCS output current achieving an output impedance of 7MΩ at 1 MHz . Another approach to increase the bandwidth and reduce the common mode of EHCS topologies is the usage of symmetric current injections. Sirtoli et al. presented a symmetric VCCS achieving a constant 1MΩ output impedance up to 300 kHz. Also, the output impedance shortened to 150 kΩ at 1 MHz . The reduced performance at higher frequencies can be also challenged by more complex designs, like the general impedance converter.
Instead of a current source, a tetra-polar BIS measurement can also be performed with a “voltage-controlled voltage source” (VCVS), if the resulting current is measured and used for the impedance calculation. A VS does not limit the maximum output current and does not limit itself to a maximum allowable current by design. Further modifications or actions have to be taken to ensure patients safety. Qureshi et al. presented a symmetric VCVS, which adresses both issues with an output impedance ranging from 7.2Ωto 13.2Ωup to 20MHz .
This work compares a modified EHCS from
Furthermore, the complexity, benchmarks and tolerances of both sources were chosen similar to allow a fair comparison.
A typical BIS measurement system consists of a front-end, a data acquisition unit and a data processing unit. The front-end generates the excitation signal, typically by a VCCS, and measures the resulting voltage. Measurements are mainly performed in the beta-dispersion range from 1 kHz to 1 MHz and should have a measurement accuracy of 0.1% or higher . The output impedance of a source determines the stability of the generated signal with respect to the connected load. To achieve a measurement accuracy of 0.1% for loads between 100 and 10 kΩ, a minimum output impedance of at least 1MΩin the mentioned frequency band is necessary. To obtain the same accuracy for a voltage source (VS), the output impedance should be at a maximum of 1.
For safe and continuous operation on biological tissue, the standard EN 60601-1-1 defines the following maximum allowable currents
In addition, the common mode current has to be below 10
As mentioned before, in BIS most commonly the EHCS is employed, along with current mirror circuits and multiple feedback operational amplifier circuitry . Theoretically, an EHCS can achieve a very high output impedance
The basic functionality of the circuitry, depicted in Figure 1, is an EHCS consisting of
we can apply the traditional balancing conditions of the EHCS as given in  to achieve a high
We note that the advantage of the circuitry as depicted in Figure 1 is the symmetric current injection from a fully differential amplifier (OP1).
The load is driven from
Other drawbacks of the traditional EHCS include instabilities due to the positive and negative feedback. Such oscillations may occur only at high frequencies and can be compensated by additional capacitors in parallel with
Because of the connection of
the identical voltage drop over both current generating resistors
According to eq. (5), large values of
In case of
The output impedance is dominated by the open loop gain value of OP1, By choosing
The output voltage swing or the saturation voltage
Thus, the theoretical maximum operable value of
The maximum load increases for smaller values of
In our application we have chosen the fully differential amplifier THS4151 (Texas Instruments, Dallas, USA) to generate the symmetrical output signal. The THS4151 has an open-loop gain
The maximum operable load can be used to determine
The basis of the VS, used in this work, is a non-inverting amplifier with a current sensing resistor
At first, the capacitor
Under the approximation
to drive a minimal load
The single-ended VS can be extended to a symmetric source shown in Figure 4, The two outputs of the single-ended VS are connected to the load. The symmetric input signal is generated by the fully differential amplifier OA2 (THS4151). The output signal and thus the dynamic range and the output impedance are doubled .
We note that the symmetric VS in Figure 4 uses two separate OAs for the generation of the symmetric output signal. A different approach from Pliquett et al.  uses only one differential OA for the final stage. The main advantage is that the negative output would be generated by the differential OA itself. This could lead to more
symmetric output signals. However, the design from Pliquett uses additional OA in the feedback path which results in a comparable amount of active components. Future work could focus on the reduction of components by omitting the active feedback OAs, which lead to a more miniaturized realization.
The operating point is a load current of 2.93mA across a 1 kΩresistor. Thus, the source should achieve an output voltage of 2.93 V over the entire frequency range and a preferably low output impedance. The THS4631 operational amplifier is used for OA1 and has a saturation voltage of
The fully differential amplifier OA2 uses a resistor ratio of 1.3 with
As both sources are symmetrical, the output signal consists of two alternating signals (
Eq. (16) describes the common mode signal
In an ideal case (
According to eq. (16),
maximum load to minimize the influence on the injected current. Thus,
For the evaluation, different offset compensation strategies can be realized by the switch
Off: No compensation with
Active & Static:
The static offset compensation adds a constant offset voltage
The output current, output impedance, harmonic distortions and common mode rejection were measured with the setup depicted in Figure 6, All signals were sampled by a Tektronix MSO2024 along with a passive test probe (1MΩ
voltage drop of two different loads (
A similar principle applies for the VS (Figure 6, right), where the output voltage with open clamps
Impedance measurements with a Keysight LCR meter E4980A yielded
The conducted research is not related to either human or animal use.
The output impedance of both sources has been optimized with respect to the bandwidth requirements. Therefore,
First, we consider the output impedance of the current source without the influence of the capacitance
achieved an output impedance of 2.15MΩ, which is 150 kΩbelow the theoretical value. Second, as this is only valid for ideal resistor conditions, we evaluated the mean output impedance for Gaussian resistor value distributions with tolerances of 0.1% (Figure 7). Here, the simulated output impedance of the EHCS is 230 k, For
Lastly, adding the capacitor
an interval of 0.6 pF. For the evaluation in hardware, this value will be also influenced by the amplifiers input- and wire-capacitances on the PCB. Thus, a trim-capacitor will be used to optimize the behavior of the EHCS in the hardware realisation.
Qureshi et al. proposed a capacitance (here:
the output impedance stays below 1.05, However, such high values of
The output impedances were calculated from the measured voltage differences across the loads
By the addition of
a significant higher value for
The theoretical maximum load of the proposed current source is
phase of the output current
In simulations, the bandwidth of the VS’s output impedance increased for higher values of
Up to 60 kHz,
In order to measure the critical lower load range for the VS, the output voltages were measured across 270 , 470Ω and 740, The measurement results are shown in Figure 12, The theoretical minimum value
The common mode rejection (CMR) circuit from Figure 5 enables us to compare the resulting common mode using no compensation, static offset correction and active compensation with the proposed feedback loop. The simulated CMR of all compensation strategies, presented in Table 1, has a nearly constant value between 1 kHz and 1 MHz, due to the low variance
The mean CMR without, with static and active compensation methods. The variances
|Compensation||[dB] ||[dB] |
|Active & Static||-128.93||0.5||-139.76||10.91|
The passive method with an
The common mode rejection was measured at 1 kΩload impedance for the three presented compensation methods. Without any compensation method, the EHCS achieved a CMR of almost -30 dB, shown in Figure 13 (blue). The CMR of the EHCS reduces from -72 dB to -43 dB with higher frequencies for the static compensation method. In contrast, the active compensation method has a fairly low frequency-dependent behavior and achieved a mean common mode reduction of -46.5 dB. The combined compensation has a mean CMR of -57.7 dB with a slightly lower reduction value for higher frequencies.
The VS showed a similar behavior as the EHCS without any compensation. The static common mode reduction of the VS also has a frequency-dependency (Figure 14). In contrast to the EHCS, the highest reduction of -80 dB is achieved for 1 kHz and 1 MHz and increases to -56.5 dB at 10 kHz. The active compensation method slightly improves for higher frequencies from -65.2 dB to -72.8 dB. Both compensation methods combined achieved a mean CMR of -71.8 dB, which is slightly below the active compensation for lower frequencies and above for higher frequencies.
The signal quality is also dependent on the spectral components aside from the measurement frequency. The relation between the signal energy at a specific measurement frequency and the energy of the remaining frequency bandwidth is called signal-to-noise ratio (SNR). The SNR of the source limits the dynamic range of the resulting bioimpedance measurements. Thus, we evaluated the resulting SNR of both sources at 10 kHz and 100 kHz. The current was measured through a 1 kΩ load with a NI-USB 6259 DAQ board (National Instruments, Austin, USA). The EHCS and the VS achieved an SNR of approx. 47 dB and 46 dB, respectively (Table 2).
SNR and THD of the hardware realization.
|SNR [dB]||10||47,53 dB||46,22 dB|
|100||47,14 dB||46,25 dB|
Besides, the sinusoidal shape of the current is another aspect of signal quality. Due to the finite slew rate of operational amplifier, the injected sinusoidal current is biased by higher harmonics of the base frequency, resulting in a deformation of the signal shape. This behavior is
quantified by the total harmonic distortion (THD) and the harmonic distortion (HD):
Measurements showed that the VS had a 1.85 dBc lower THD compared to the EHCS at 10 kHz. This effect diminished at 100 kHz, where both sources achieved a THD of less then -35 dBc.
A more detailed investigation of the harmonic distortion has been measured with the Tektronix MSO2024, The second harmonic distortions (HD2, solid lines) of the EHCS and the VS are constant at -68 dBc up to 100 kHz, which is equal to the noise floor of the oscilloscope (Figure 15). Above 100 kHz, HD2 of the VS rises constantly with 20 dBc per decade. For the EHCS, HD2 rises at 300 kHZ with 60 dBc per decade. Both sources reach approx. -30 dBc at 1 MHz. Typical for differential signals, the third harmonics of both sources (HD3, dashed lines) are above the second harmonics. On average, the EHCS had a HD3 of -50 dBc and the VS has -53 dBc. The third order harmonics shows no frequency-dependent behavior in contrast to the second order harmonics. The values of the fourth as well as the fifth harmonic distortion are below the noise floor of the used oscilloscope (-64 dB).
The parallel capacitors
For the VS,
The acceptable load of the EHCS has good characteristics in the lower load range. The difficulties of the EHCS occur at loads above 2.47 kΩ at higher frequencies (above 500 kHz). Such large bioimpedances do not usually occur at these frequencies, hence this impairment is negligible. The voltage source exhibits phase shifts and distortions above 470, Yet, amplitude attenuation is not affected.
The common mode signals of voltage and current sources are influenced by offset voltages of the operational amplifiers and miss-matches of resistors. Different common mode reduction strategies were evaluated. We note that the measured common mode of the EHCS might be influenced by the probe loading effect especially for the high impedance positive output. The usage of an instrumentation amplifier, similar to the measurement setup of the output impedance, is not applicable due to the common mode measurement principle. However, the probe loading effect applies to all compensation strategies and should not compromise the comparability. We could significantly reduce the common mode of both sources in simulations as well as in experimental evaluation by the application of a static offset compensation. Additionally, an active feedback compensation could also reduce the common mode signal, but to a minor extent. However, the common mode signal increases for higher frequencies. This behavior can be reduced by the usage of active compensation and results in a more constant common mode reduction over the whole frequency range. The combination of both methods achieved a mean common mode reduction by -71.8 dB for the voltage source, which is similar to the passive compensation but shows significantly less frequency-dependency. For the EHCS, the combined usage of both methods is equivalent to the purely static compensation with -57.7 dB, but has a more constant frequency behavior. According to the EN 60601-1, a maximum current of 10
For the signals at 10 kHz and 100 kHz, the SNRs are constant and almost identical with approximately 47 dB. The THD, on the other hand, increases from -42.3 dBc by 7.1 dBc for the EHCS, and from -44.2 dBc by 8.4 dBc for the VS. Both sources show the same frequency-dependent behavior, which might be caused by the THD of the differential amplifier THS4151.
The usage of symmetric signals suppresses HDs. The measured HD2 were below the noise floor of -68 dBc for both sources up to 100 kHz. For higher frequencies, HD2 increased, which might be caused by higher frequency poles of the sources.
The mean CMR without, with static and active compensation methods. The variances σ of the VS (Cv = 82 pF) and current source (Cc = 2.6 pF) are given for the frequency range 1 kHz to 1 MHz.
|Active & Static||-128.93||0.5||-139.76||10.91|
SNR and THD of the hardware realization.
|SNR [dB]||10||47,53 dB||46,22 dB|
|100||47,14 dB||46,25 dB|