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Simulation Research of Electrostatic Precipitator Power Supply Voltage Control System Based on Finite Element Differential Equation

Publicado en línea: 15 Jul 2022
Volumen & Edición: AHEAD OF PRINT
Páginas: -
Recibido: 14 Feb 2022
Aceptado: 13 Apr 2022
Detalles de la revista
License
Formato
Revista
eISSN
2444-8656
Primera edición
01 Jan 2016
Calendario de la edición
2 veces al año
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Inglés
Introduction

At present, electrostatic dust removal equipment is an internationally recognized high-efficiency dust removal device. The high-voltage power supply system is the leading electrical part of the device. Its output voltage directly affects the working effect of the electrostatic precipitator. Generally, high-voltage power supplies for electrostatic precipitators can be divided into three categories: power frequency power supplies, high-frequency power supplies, and pulsed high-voltage power supplies. Due to its low frequency, the power frequency electrostatic precipitator power supply has low power density, significant voltage fluctuation, and low dust removal efficiency. It cannot adapt to the working conditions of high concentration dust and high specific resistance dust. High-frequency power supplies have been improved for this problem [1]. The method adopts high-frequency control to achieve the goals of high power density, small voltage ripple, and high dust removal efficiency. However, it has not been widely used due to its stability and other reasons. Both the power frequency power supply and the high-frequency power supply belong to the high voltage DC power supply.

From the principle of electrostatic precipitator, the pulse power supply is more suitable. It has the following advantages: the pulse voltage is short and less prone to flashover [2]. The high pulse voltage peak value increases the dust charge and improves dust removal efficiency. The pulse power supply is more beneficial for collecting high specific resistance dust. It can suppress the back corona phenomenon and improve the dust removal efficiency.

The development of electrostatic precipitator pulse power supply has formed two topologies of high-voltage side pulse and low-voltage side pulse. The load of the electrostatic precipitator varies greatly, and the pulse power supply has pulse tailing on the pack. This is a complex problem in the design of pulsed power systems. In this paper, the modal analysis is carried out for the high voltage topology of the pulsed power supply, and the differential equations in different modalities are obtained [3]. This paper conveys the effects of parameters such as the coupling inductance, coupling coefficient, and parasitic resistance of the dust chamber on the load oscillation by deriving the transfer function and three-dimensional drawing. At the same time, it provides an essential basis for designing a pulse power supply. Finally, this paper proves the correctness and feasibility of theoretical analysis and design through simulation and experiment.

Working modal analysis of electrostatic precipitator pulse power supply
Circuit Decoupling

The circuit topology of the high-voltage side of the electrostatic precipitator pulse power supply is shown in Figure 1. The circuit is divided into three parts: pulse power supply module, resonant cavity, and DC power supply module.

Figure 1

High Side Circuit Topology

The pulse power supply module consists of a DC power supply VPS and a coupled inductor LPS. Its principal function is to provide pulse voltage [4]. The resonant cavity is vibrant inductor L, high voltage switch SW, precipitator load, and coupling capacitor C. Among them, the high-voltage switch is composed of a thyristor anti-parallel diode. The precipitator load is equivalent to capacitance Cf and variable resistance Rf. The primary function of the resonant cavity is to couple the DC supply voltage and the pulsed supply voltage. We utilize narrow pulse high-pressure dust removal. The DC power supply module is composed of a DC power supply VDC and a coupled inductor LDC. The coupling inductance LPS and the coupling inductance LDC are connected through the magnetic core. The function of the DC power supply module is to increase the DC voltage [5]. Dual power supplies to power the circuit with coupled inductors between the power supplies. This brings difficulties to the analysis work, and it is not easy to obtain an intuitive circuit mesh. So we move the coupled inductor and decouple it to get Figure 2. The equivalent circuit diagram removes the coupled inductance to acquire three new inductances: mutual inductance M, pulse-side even inductance Lσ1 and DC-side leakage inductance Lσ2. The quantitative relationship is shown in formulas (1) to (2):

Figure 2

Circuit Decoupling Equivalent

LPS=LDC=LCP {L_{PS}} = {L_{DC}} = {L_{CP}} M=kLCP M = k{L_{CP}} Lσ1=Lσ2=(1k)LCP {L_{\sigma 1}} = {L_{\sigma 2}} = (1 - k){L_{CP}}

Modal analysis of the circuit

Figure 2 shows the equivalent decoupling circuit. After a steady-state, the course can be comparable to three meshes. Where i1, i2, i3 is the current of the three meshes. After the thyristor is turned on, the inductor L and the load capacitor Cf form resonance. i1 first flows through the thyristor to zero and then through the diode to turn off the thyristor. The diode freewheels to zero, and the resonance process ends. During the resonance process, the current i1 changes in a sine wave. The voltage on the load capacitor is a DC superimposed cosine wave voltage [6]. The mesh current i2, i3 also changes during the resonance process, but due to the sizeable mutual inductance M, the difference between i2, i3 and i1 is two orders of magnitude. Therefore, its influence is ignored in the resonance process.

Figure 3 shows the oscillation mode after the end of the resonance. After the resonance is over, the thyristor has been turned off in the freewheeling diode stage. Therefore, the branch where the resonant inductor is located is disconnected. The five elements of mutual inductance M, leakage inductance Lσ1, Lσ2, C and Cf resonate.

Figure 3

Oscillation transition after resonance

The load voltage waveform with reasonable system parameters is shown in Figure 4. The load voltage overshoot in the oscillation phase is slight, and the regulation time is short. Figure 5 shows the load voltage waveform with unreasonable parameter design. The significant fluctuation of the DC voltage causes the pulse voltage not to match the expected value. This waveform is not easy to control, seriously damaging the system [7]. Therefore, we must carry out mathematical modeling and detailed parameter design of the system.

Figure 4

Load Voltage Waveform 1

Figure 5

Load Voltage Waveform 2

Circuit mathematical model and parameter design
Circuit modal mathematical model

Mode 1 is the resonance state. We found five state variables from Fig. 2 to construct the differential equation (4) { CfddtVCf(t)+VCf(t)Rf=i1(t)i3(t)CddtVC(t)=i1(t)Lddt[ i1(t)+i2(t) ]+VCf(t)+VC(t)+R1[ i1(t)+i2(t) ]=0Lσ2ddti3(t)+M[ ddti3(t)+ddti2(t) ]=VDC+VCf(t)Lσ1ddti2(t)+M[ ddti3(t)+ddti2(t) ]=VPS+[ VCf(t)+VC(t) ] \left\{ \matrix {C_f}\frac{d}{{dt}}{V_{Cf}}(t) + \frac{{{V_{Cf}}(t)}}{{{R_f}}} = {i_1}(t) - {i_3}(t) \hfill \cr C\frac{d}{{dt}}{V_C}(t) = {i_1}(t) \hfill \cr L\frac{d}{{dt}}\left[ {{i_1}(t) + {i_2}(t)} \right] + {V_{Cf}}(t) + {V_C}(t) + {R_1}\left[ {{i_1}(t) + {i_2}(t)} \right] = 0 \hfill \cr {L_{\sigma 2}}\frac{d}{{dt}}{i_3}(t) + M\left[ {\frac{d}{{dt}}{i_3}(t) + \frac{d}{{dt}}{i_2}(t)} \right] = {V_{DC}} + {V_{Cf}}(t) \hfill \cr {L_{\sigma 1}}\frac{d}{{dt}}{i_2}(t) + M\left[ {\frac{d}{{dt}}{i_3}(t) + \frac{d}{{dt}}{i_2}(t)} \right] = - {V_{PS}} + \left[ {{V_{Cf}}(t) + {V_C}(t)} \right] \hfill \cr \endmatrix \right. { VCF(t0)=VDCVC(t0)=VPS+VDCi3(t0)=VDCRfi2(t0)=0i1(t0)=0 \left\{ \matrix {V_{CF}}({t_0}) = - {V_{DC}} \hfill \cr {V_C}({t_0}) = {V_{PS}} + {V_{DC}} \hfill \cr {i_3}({t_0}) = \frac{{{V_{DC}}}}{{{R_f}}} \hfill \cr {i_2}({t_0}) = 0 \hfill \cr {i_1}({t_0}) = 0 \hfill \cr \endmatrix \right.

In formula (4), VCf is the load capacitor voltage. VC is the coupling capacitor voltage. R1 is the resonant inductor resistance. The initial value of the resonant state circuit is equation (5). The system of differential equations (4) is a higher-order differential equation containing five unknowns. In this paper, we need to use the Laplace transform and bring in the initial value (5) to obtain the numerical solutions of i1 (t), i2 (t), i3 (t), VCf (t) and VC (t).

Mode 2 is the oscillation state: the initial value of the oscillation state circuit is the final value of the resonant state circuit. We can calculate the resonance period T1 from equations (7) to (9). We bring T1 into i2 (t), i3 (t), VCf (t) and VC (t) to get the initial oscillation values i2 (T1), i3 (T1), VCf (T1) and.

Figure 3 shows that four state variables can be established in the differential equation system (6). Oscillation state differential equation system contains four unknowns high-order differential equation system. Similarly, we need to obtain numerical solutions to use the Laplace transform and inverse transform. { CfddtVCf(t)+VCf(t)Rf=[ i3(t)+i2(t) ]CddtVC(t)=i2(t)Lσ2ddti3(t)+M[ ddti3(t)+ddti2(t) ]=VDC+VCf(t)Lσ1ddti2(t)+M[ ddti3(t)+ddti2(t) ]=VPS+[ VCf(t)+VC(t) ] \left\{ \matrix {C_f}\frac{d}{{dt}}{V_{Cf}}(t) + \frac{{{V_{Cf}}(t)}}{{{R_f}}} = - \left[ {{i_3}(t) + {i_2}(t)} \right] \hfill \cr C\frac{d}{{dt}}{V_C}(t) = - {i_2}(t) \hfill \cr {L_{\sigma 2}}\frac{d}{{dt}}{i_3}(t) + M\left[ {\frac{d}{{dt}}{i_3}(t) + \frac{d}{{dt}}{i_2}(t)} \right] = {V_{DC}} + {V_{Cf}}(t) \hfill \cr {L_{\sigma 1}}\frac{d}{{dt}}{i_2}(t) + M\left[ {\frac{d}{{dt}}{i_3}(t) + \frac{d}{{dt}}{i_2}(t)} \right] = - {V_{PS}} + \left[ {{V_{Cf}}(t) + {V_C}(t)} \right] \hfill \cr \endmatrix \right. Cx=CCfC+Cf {C_x} = \frac{{C\,{C_f}}}{{C + {C_f}}} ω1=1LCx {\omega _1} = \frac{1}{{\sqrt {L\,{C_x}} }} T1=2πω1 {T_1} = \frac{{2\pi }}{{{\omega _1}}}

Parameter design of pulse power supply

The load current and voltage in the resonant state are mainly related to Cf and L. The load voltage in the oscillating state is primarily associated with the coupling inductance LCP, the coupling coefficient k, the resistance value R1 of the resonant inductance, and the initial value of Sinian. Since the oscillation state contains five energy storage elements, the numerical analysis method analyzes the system parameters [8]. The specific parameters are shown in Table 1.

System Parameters

Parameter name Numerical value
Pulse side voltage VPS/V 380
DC side voltage VDC/V 790
Load resistance Rf/Ω 950
Load capacitance Cf/nF 100
Resonant inductance L/μH 101.3
Coupling capacitor C/μF 1
Resonance period T1/μs 20
Resonant inductor resistance R1/Ω 0.6

Cf is the equivalent capacitance between the electrodes. We control it within the range of 5000pF to 100000pF. The load current of the dust collector is generally 800 ~ 1200mA. In this paper, Cf=100nF, and the load current is 0.8A, so Rf =950Ω. The coupling capacitor C is usually 5 to 10 times the load capacitance, C=1μF.

Influence of resonant inductor resistance R1

From the analysis of mode 1, it can be known that the more significant R1 is, the more resonance energy loss is. At this time, the difference between the final value of the resonance voltage of VCf and VC and the initial value is more significant. This also means that the initial value of the second-stage shock is also more enormous. This resulted in increased oscillation voltage overshoot and increased regulation time. This isn’t good for the system [9]. Therefore, the resonant inductor resistance should be minimized. In the subsequent calculation, we used the inductance resistance value of the experimental method of 0.6.

Optimal design of LCP and k

The design of the coupled inductor LCP directly affects the oscillation state. The oscillating state contains five energy storage elements. They all participate in resonance. In this paper, the decomposition analysis of each part of the excitation response to the load voltage after resonance is carried out. FIG. 6 is a circuit diagram of the Laplace transform in an oscillating state. We use formulas (1) to (9) to calculate the final value of the resonance state. We bring it into the excitation in Figure 6 and multiply it by the corresponding transfer function to obtain each part of the excitation response to the load voltage, as shown in Figure 7.

Figure 6

Laplace circuit diagram in the oscillation state

Figure 7

Load Capacitor Voltage Response to Load

It can be seen from Fig. 7 that the oscillation state can be divided into oscillation 1 and oscillation 2. Oscillation 1 determines the overshoot of the load voltage, which is determined by the load voltage excitation FCfV (t), the mutual inductance excitation FMV (t), and the DC side dual inductance excitation lσ2V (t). Oscillation 2 defines the adjustment time of the load oscillation [10]. The coupling capacitor excitation mainly determines it FMV (t). CCfV=s2RfCf[ s2(2LσMC+L σ2C)+Lcp ](s2RfCf+s)[ s2(2LσMC+L σ2C)+Lcp ]+Rf(1+2s2LσC) {C_{CfV}} = \frac{{{s^2}{R_f}{C_f}\left[ {{s^2}\left( {2{L_\sigma }MC + L_\sigma ^2C} \right) + {L_{cp}}} \right]}}{{\left( {{s^2}{R_f}{C_f} + s} \right)\left[ {{s^2}\left( {2{L_\sigma }MC + L_\sigma ^2C} \right) + {L_{cp}}} \right] + {R_f}\left( {1 + 2{s^2}{L_\sigma }C} \right)}} CCV=s2LσC[ s2MCfRf+sM+Rf ]s2LσC[ s2MCfRf+sM+Rf ]+(1+s2LσC)(s2LcpCfRf+sLcp+Rf)RfRf+sM(1+sCfRf) {C_{CV}} = \frac{{{s^2}{L_\sigma }C\left[ {{s^2}M{C_f}{R_f} + sM + {R_f}} \right]}}{{{s^2}{L_\sigma }C\left[ {{s^2}M{C_f}{R_f} + sM + {R_f}} \right] + \left( {1 + {s^2}{L_\sigma }C} \right)\left( {{s^2}{L_{cp}}{C_f}{R_f} + s{L_{cp}} + {R_f}} \right)}}\frac{{{R_f}}}{{{R_f} + sM(1 + s{C_f}{R_f})}} GMV=(1+s2LσC)sRfLσ(1+s2LσC)sLσ+(1+2s2LσC)[ Rf+sM(1+sCfRf) ] {G_{MV}} = \frac{{\left( {1 + {s^2}{L_\sigma }C} \right)s{R_f}{L_\sigma }}}{{\left( {1 + {s^2}{L_\sigma }C} \right)s{L_\sigma } + \left( {1 + 2{s^2}{L_\sigma }C} \right)\left[ {{R_f} + sM\left( {1 + s{C_f}{R_f}} \right)} \right]}} CLσ1V=(1+s2LσC)[ s2MCfRf+sM+Rf ]sRfCLσ+(1+s2LσC)(1+sRfCf)Lσ+(1+s2LσC)(s2MCfRf+sM+Rf)RfRf+sM(1+sCfRf) {C_{{L_{\sigma 1}}V}} = \frac{{\left( {1 + {s^2}{L_\sigma }C} \right)\left[ {{s^2}M{C_f}{R_f} + sM + {R_f}} \right]}}{{s{R_f}C{L_\sigma } + \left( {1 + {s^2}{L_\sigma }C} \right)\left( {1 + s{R_f}{C_f}} \right){L_\sigma } + \left( {1 + {s^2}{L_\sigma }C} \right)\left( {{s^2}M{C_f}{R_f} + sM + {R_f}} \right)}}\frac{{{R_f}}}{{{R_f} + sM(1 + s{C_f}{R_f})}} CLσ2V=s2LσC[ s2MCfRf+sM+Rf ]s2LσC[ s2MCfRf+sM+Rf ]+(1+s2LσC)(s2LcpCfRf+sLcp+Rf)RfRf+sM(1+sCfRf) {C_{{L_{\sigma 2}}V}} = \frac{{{s^2}{L_\sigma }C\left[ {{s^2}M{C_f}{R_f} + sM + {R_f}} \right]}}{{{s^2}{L_\sigma }C\left[ {{s^2}M{C_f}{R_f} + sM + {R_f}} \right] + \left( {1 + {s^2}{L_\sigma }C} \right)\left( {{s^2}{L_{cp}}{C_f}{R_f} + s{L_{cp}} + {R_f}} \right)}}\frac{{{R_f}}}{{{R_f} + sM(1 + s{C_f}{R_f})}}

We extract the parts of FCfV (t), FMV (t) and Lσ2V (t) with the same attenuation coefficient to constitute oscillation 1. We can simplify the description by equations (15)~(22): Vz1=P1eα1tcos(ω2t)P1α1ω1sin(ω2t)eα1t+P2CfLcpω2sin(ω3t)eα1t+P3CfMω3sin(ω4t)eα1t \eqalign{ & {V_{z1}} = {P_1}{e^{ - {\alpha _1}t}}\cos \left( {{\omega _2}t} \right) - \frac{{{P_1}\,{\alpha _1}}}{{{\omega _1}}}\sin \left( {{\omega _2}t} \right){e^{ - {\alpha _1}t}} + \cr & \frac{{{P_2}}}{{{C_f}{L_{cp}}{\omega _2}}}\sin \left( {{\omega _3}t} \right){e^{ - {\alpha _1}t}} + \frac{{{P_3}}}{{{C_f}M{\omega _3}}}\sin ({\omega _4}t){e^{ - {\alpha _1}t}} \cr} ω2=2Cf(2M+Lσ)14(RfCf)2 {\omega _2} = \sqrt {\frac{2}{{{C_f}(2M + {L_\sigma })}} - \frac{1}{{4{{\left( {{R_f}{C_f}} \right)}^2}}}} ω3=1Cf(M+Lσ)14(RfCf)2 {\omega _3} = \sqrt {\frac{1}{{{C_f}(M + {L_\sigma })}} - \frac{1}{{4{{\left( {{R_f}{C_f}} \right)}^2}}}} ω4=1CfM14RfCf2 {\omega _4} = \sqrt {\frac{1}{{{C_f}M}} - \frac{1}{{4{R_f}\mathop {{C_f}}\nolimits^2 }}} α1=12RfCf {\alpha _1} = \frac{1}{{2{R_f}{C_f}}} P1=VDCVCf(T1) {P_1} = {V_{DC}} - {V_{Cf}}({T_1}) P2=iM {P_2} = i\,\,M P3=(i3VdcRf)Lσ {P_3} = \left( {{i_3} - \frac{{{V_{dc}}}}{{{R_f}}}} \right){L_\sigma }

Experimental waveforms

The experimental parameters are shown in Table 1. The observed waveform is shown in Figure 8. LCP is 310 m, k is 0.75. The load voltage superimposes a cosine pulse based on the DC negative voltage [11]. The resonant period is 20 μs. The pulse voltage is close to 2VPS. The pulse current flows through the coupling capacitor during the resonance phase, resulting in a voltage gap. The voltage gap value is around 75V, and the coupling capacitor voltage is stable at 1140V. i2 represents the DC side pulse supply current. The resonance phase i2 is increased by 0.2A. This is more than two orders of magnitude different from the 15A of the resonant current. This also shows that the system parameter design is more reasonable.

Figure 8

Experimental waveforms

Conclusion

In this paper, the modal analysis of the circuit on the high-voltage side of the electrostatic precipitator power supply is carried out, and its mathematical model is constructed. We obtain the final resonance value of each part according to the Laplace transform circuit. The excitation of coupled inductor voltage and pulse-side leakage inductor current affects the load voltage regulation time.

Figure 1

High Side Circuit Topology
High Side Circuit Topology

Figure 2

Circuit Decoupling Equivalent
Circuit Decoupling Equivalent

Figure 3

Oscillation transition after resonance
Oscillation transition after resonance

Figure 4

Load Voltage Waveform 1
Load Voltage Waveform 1

Figure 5

Load Voltage Waveform 2
Load Voltage Waveform 2

Figure 6

Laplace circuit diagram in the oscillation state
Laplace circuit diagram in the oscillation state

Figure 7

Load Capacitor Voltage Response to Load
Load Capacitor Voltage Response to Load

Figure 8

Experimental waveforms
Experimental waveforms

System Parameters

Parameter name Numerical value
Pulse side voltage VPS/V 380
DC side voltage VDC/V 790
Load resistance Rf/Ω 950
Load capacitance Cf/nF 100
Resonant inductance L/μH 101.3
Coupling capacitor C/μF 1
Resonance period T1/μs 20
Resonant inductor resistance R1/Ω 0.6

Preger, C., Overgaard, N. C., Messing, M. E., & Magnusson, M. H. Predicting the deposition spot radius and the nanoparticle concentration distribution in an electrostatic precipitator. Aerosol Science and Technology.,2020; 54(6):718–728PregerCOvergaardNCMessingMEMagnussonMHPredicting the deposition spot radius and the nanoparticle concentration distribution in an electrostatic precipitatorAerosol Science and Technology202054671872810.1080/02786826.2020.1716939Search in Google Scholar

Zhu, Y., Chen, C., Shi, J., & Shangguan, W. Enhancement of air purification by unique W-plate structure in two-stage electrostatic precipitator: A novel design for efficient capture of fine particles. Advanced Powder Technology.,2020; 31(4):1643–1658ZhuYChenCShiJShangguanW.Enhancement of air purification by unique W-plate structure in two-stage electrostatic precipitator: A novel design for efficient capture of fine particlesAdvanced Powder Technology20203141643165810.1016/j.apt.2020.02.003Search in Google Scholar

Feng, Y., Gao, W., Zhou, M., Luo, K., Fan, J., Zheng, C., & Gao, X. Numerical modeling on simultaneous removal of mercury and particulate matter within an electrostatic precipitator. Advanced Powder Technology.,2020; 31(4):1759–1770FengYGaoWZhouMLuoKFanJZhengCGaoX.Numerical modeling on simultaneous removal of mercury and particulate matter within an electrostatic precipitatorAdvanced Powder Technology20203141759177010.1016/j.apt.2020.01.037Search in Google Scholar

Oualid, I., Flazi, S., Oussalah, N., Naoui, N., Benamar, H., & Stambouli, A. B. Numerical Simulation of Polluting Particles’ Trajectory Inside an Electrostatic Precipitator of Multi-Wire-to-Plate Electrodes. Elektrotehniski Vestnik.,2021; 88(5):255–266OualidIFlaziSOussalahNNaouiNBenamarHStambouliABNumerical Simulation of Polluting Particles’ Trajectory Inside an Electrostatic Precipitator of Multi-Wire-to-Plate ElectrodesElektrotehniski Vestnik2021885255266Search in Google Scholar

Sadeghpour, A., Oroumiyeh, F., Zhu, Y., Ko, D. D., Ji, H., Bertozzi, A. L., & Ju, Y. S. Experimental study of a string-based counterflow wet electrostatic precipitator for collection of fine and ultrafine particles. Journal of the Air & Waste Management Association.,2021; 71(7):851–865SadeghpourAOroumiyehFZhuYKoDDJiHBertozziALJuYSExperimental study of a string-based counterflow wet electrostatic precipitator for collection of fine and ultrafine particlesJournal of the Air & Waste Management Association202171785186510.1080/10962247.2020.186962733395565Search in Google Scholar

Gençoğlu, M. & Agarwal, P. Use of Quantum Differential Equations in Sonic Processes. Applied Mathematics and Nonlinear Sciences.,2021; 6(1): 21–28GençoğluMAgarwalP.Use of Quantum Differential Equations in Sonic ProcessesApplied Mathematics and Nonlinear Sciences202161 212810.2478/amns.2020.2.00003Search in Google Scholar

Wu, M., Payshanbiev, A., Zhao, Q. & Yang, W. Nonlinear optimization generating the Tomb Mural Blocks by GANS. Applied Mathematics and Nonlinear Sciences., 2021;6(1): 43–56WuMPayshanbievAZhaoQ.YangW.Nonlinear optimization generating the Tomb Mural Blocks by GANSApplied Mathematics and Nonlinear Sciences202161435610.2478/amns.2020.2.00072Search in Google Scholar

Sung, J. H., Kim, S., Kim, S., Han, B., Kim, Y. J., & Kim, H. J. Development of an Integrated Electrostatic Precipitator and Wet Scrubber System for Controlling No x and Particulate Matter Emissions from a Semiconductor Manufacturing Process. IEEE Transactions on Industry Applications.,2020; 56(6):7012–7019SungJHKimSKimSHanBKimYJKimHJDevelopment of an Integrated Electrostatic Precipitator and Wet Scrubber System for Controlling No x and Particulate Matter Emissions from a Semiconductor Manufacturing ProcessIEEE Transactions on Industry Applications20205667012701910.1109/TIA.2020.3023670Search in Google Scholar

Tahir, M. S., Gruber, C., Siebenhofer, M., & Saleem, M. Impact of the design of discharge electrode on the current/voltage characteristics and the rate of migration in electrostatic precipitation (ESP). The Nucleus.,2020; 46(3):231–235TahirMSGruberCSiebenhoferMSaleemM.Impact of the design of discharge electrode on the current/voltage characteristics and the rate of migration in electrostatic precipitation (ESP)The Nucleus2020463231235Search in Google Scholar

Steefel, C. I., & Tournassat, C. A model for discrete fracture-clay rock interaction incorporating electrostatic effects on transport. Computational Geosciences., 2021;25(1):395–410SteefelCITournassatCA model for discrete fracture-clay rock interaction incorporating electrostatic effects on transportComputational Geosciences202125139541010.1007/s10596-020-10012-3Search in Google Scholar

Zhang, Y., Wang, X., Zhang, H., Liu, J., & Luan, T. Numerical simulation of WFGD wastewater with atomizing and crystallization treatment. Energy Sources, Part A: Recovery, Utilization, and Environmental Effects.,2020; 42(10):1268–1285ZhangYWangXZhangHLiuJLuanT.Numerical simulation of WFGD wastewater with atomizing and crystallization treatmentEnergy Sources, Part A: Recovery, Utilization, and Environmental Effects202042101268128510.1080/15567036.2019.1604852Search in Google Scholar

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